Signal converting apparatus and receiving apparatus for supporting concurrent dual bands in wireless communication system

ABSTRACT

A receiving apparatus in a wireless communication system includes: an antenna configured to receive a wireless frequency signal including a first frequency band signal and a second frequency band signal; a low noise amplifier (LNA) configured to amplify the wireless frequency signal, output the first frequency band signal as a differential phase signal, and output the second frequency band signal as a common phase signal; a differentiator configured to pass only the differential phase signal between the signals outputted from the LNA; and a combiner configured to pass only the common phase signal between the signals outputted from the LNA.

BACKGROUND OF THE INVENTION

The present invention relates to a signal converting apparatus and areceiving apparatus for supporting concurrent dual bands in a wirelesscommunication system.

Currently, the wireless communication employs technology to increase adata rate using a wideband or multiband, and the research has beenactively conducted on a radio frequency (RF) receiver supporting thetechnology. Such an RF receiver may increase the flexibility of chips tothereby reduce the unit cost. However, the RF receiver has adisadvantage in that the quality of the wireless communication decreasesdue to the interference between frequency bands corresponding towireless communication application fields. In order to support wirelesscommunication at a high data rate while minimizing the interference, thepower consumption and chip size of the receiver should be increased. Inthis case, the competitiveness of the receiver may be reduced.

An ultra-wide band intellectual RF receiver capable of processing a widefrequency band of several GHz may obtain an effect of implementingvarious systems at a low cost. However, since the power consumption ofthe receiver is high and several frequency bands are amplified by thesame gain and then applied to a system, the interference between signalsinevitably increases. Meanwhile, a multiband intellectual RF receivercapable of realizing a narrow-band characteristic in two or morefrequency bands deals with only a requested frequency band. Therefore,the power consumption of the receiver is low. Even at this time,however, an interference problem still exists. In this case, when aplurality of RF receivers dealing with the respective frequency bandsare used, the interference may be minimized. However, the chip size andpower consumption thereof are increased, which makes it impossible tomaximize the price competitiveness.

SUMMARY OF THE INVENTION

An embodiment of the present invention is directed to an RF receivercapable of minimizing a chip size and power consumption while minimizinginterference between frequency bands in a dual-band system.

In accordance with an embodiment of the present invention, a receivingapparatus in a wireless communication system includes: an antennaconfigured to receive a wireless frequency signal including a firstfrequency band signal and a second frequency band signal; a low noiseamplifier (LNA) configured to amplify the wireless frequency signal,output the first frequency band signal as a differential phase signal,and output the second frequency band signal as a common phase signal; adifferentiator configured to pass only the differential phase signalbetween the signals outputted from the LNA; and a combiner configured topass only the common phase signal between the signals outputted from theLNA.

In accordance with another embodiment of the present invention, there isprovided a load circuit of an LNA, including a first capacitor, a secondcapacitor, a third capacitor, a first inductor, and a second inductor.The third capacitor is coupled between a first node and a second node,one ends of the first inductor and the first capacitor are coupledbetween the first node and the third capacitor, the second inductor andthe second capacitor are coupled between the second node and the thirdcapacitor, and the other ends of the first inductor, the firstcapacitor, the second inductor, and the second capacitor are coupled toa ground terminal.

In accordance with another embodiment of the present invention, an LNAincludes an input terminal and a first intermediate tap inductorincluding: an impedance matching unit configured to match a single-phasesignal inputted through the input terminal; first and second inductorscoupled to the impedance matching unit and having one ends coupledsymmetrically through a first tap; and a first second parasiticcapacitor and a second capacitor being symmetrical with the first tap,wherein the first intermediate tap inductor configured to convert thesingle-phase signal outputted from the impedance matching unit into adifferential phase signal and a common phase signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram illustrating a receiving apparatus in a wirelesscommunication system according to the embodiment of the presentinvention.

FIG. 2 is a circuit diagram illustrating a load unit of a signalconverting apparatus according to a first embodiment of the presentinvention.

FIG. 3 is a graph showing a simulation result for transfer functionvalues at nodes P1 and P2 in the circuit diagram of FIG. 2.

FIG. 4 is a graph showing phases at the nodes P1 and P2 in the circuitdiagram of FIG. 2.

FIG. 5 is a graph showing a phase difference between the node P1 and thenode P2 in the circuit diagram of FIG. 2.

FIG. 6 is a circuit diagram illustrating a load unit of a signalconverting apparatus according to a second embodiment of the presentinvention.

FIG. 7 is a graph showing a simulation result for transfer functionvalues at the nodes P1 and P2 in the circuit diagram of FIG. 6.

FIG. 8 is a diagram illustrating a load unit of a signal convertingapparatus according to a third embodiment of the present invention.

FIG. 9 is a graph showing a simulation result for transfer functionvalues at the nodes P1 and P2 in the circuit diagram of FIG. 8.

FIG. 10 is a circuit diagram illustrating an example in which thecircuit diagram of FIG. 2 is applied to the design of a two-stage LNA.

FIG. 11 is a graph showing gain characteristics and phase differencecharacteristics for the circuit diagram of FIG. 10.

FIG. 12 is a graph showing a transient simulation result of dual-bandoutput signals in the circuit diagram of FIG. 10.

DESCRIPTION OF SPECIFIC EMBODIMENTS

Exemplary embodiments of the present invention will be described belowin more detail with reference to the accompanying drawings. The presentinvention may, however, be embodied in different forms and should not beconstructed as limited to the embodiments set forth herein. Rather,these embodiments are provided so that this disclosure will be thoroughand complete, and will fully convey the scope of the present inventionto those skilled in the art. Throughout the disclosure, like referencenumerals refer to like parts throughout the various figures andembodiments of the present invention.

In the entire specification and claims, when a certain unit“includes/comprises” a certain component, it means that the unit mayfurther include/comprise another component without excluding anothercomponent, as long as specific descriptions are not made.

Now, a signal converting apparatus and a receiving apparatus in awireless communication system according to an embodiment of the presentinvention will be described in detail with reference to the accompanyingdrawings.

FIG. 1 is a diagram illustrating the receiving apparatus in a wirelesscommunication system according to the embodiment of the presentinvention.

Referring to FIG. 1, the receiving apparatus in a wireless communicationsystem according to the embodiment of the present invention includes anantenna 100, a low noise amplifier (LAN) 102, and a frequency downconverter 103.

The antenna 100 is configured to directly receive an RF signal from atransmitting apparatus in a wireless communication system.

The LNA 102 includes a transconductance (Gm) unit 102 a configured toconvert an input voltage into a current and a load unit 102 b configuredto convert the current into a voltage. The LNA 102 amplifies a selectedsignal while minimizing noise. The LNA 102 converts the amplified signalinto a differential phase signal and a common phase signal at the sametime for each frequency band. Accordingly, in this specification, theLNA 102 may be used together with a signal converting apparatus.

The frequency down converter 103 includes a first frequencydown-conversion switching stage 103 a, a second frequencydown-conversion switching stage 103 b, a differentiator 103 c, and acombiner 103 d.

The first frequency down-conversion switching stage 103 a is configuredto down-convert a differential signal of a first frequency band into alocal oscillator frequency corresponding to the first frequency band. Atthis time, a signal of a second frequency band is also down-convertedinto a common mode signal.

The differentiator 103 c is configured to pass only a differentialsignal of the first frequency band and offset the command mode signal ofthe second frequency band, between the down-converted dual-band signals.

The second frequency down-converting switching stage 103 b is configuredto down-convert the common mode signal of the second frequency band intoa local oscillator frequency corresponding to the second frequency band.At this time, a signal of the first frequency band is alsodown-converted.

The combiner 103 d is configured to pass only the common mode signal ofthe second frequency band and offset the differential signal of thefirst frequency, between the down-converted dual-band signals.

Here, the positions of the frequency down-conversion switching stages,the differentiator, and the combiner may be changed. Even at this time,a small area may be implemented at low power as in the above-describedconfiguration. FIG. 1 illustrates that the differentiator 103 c and thecombiner 103 d are provided separately from the frequencydown-conversion switching stages 103 a and 103 b. However, thedifferentiator 103 c and the combiner 103 d may be implemented in atransconductance form of the frequency down converter 103.

Referring to FIGS. 2 to 12, a signal converting apparatus in a wirecommunication system according to the embodiment of the presentinvention will be described in detail.

FIG. 2 illustrates a load unit 102 b of a signal converting apparatusaccording to a first embodiment of the present invention.

Referring to FIG. 2, the load unit 102 b of the signal convertingapparatus includes capacitors C1 to C3 and inductors L1 and L2. Forconvenience of description, it is assumed that the capacitors C1 to C3and the inductors L1 and L2 are ideal passive elements.

The capacitor C3 is coupled between a node P1 and a node P2, one ends ofthe capacitor C1 and the inductor L1 are coupled between the node P1 andthe capacitor C3, and one ends of the capacitor C2 and the inductor L2are coupled between the node P2 and the capacitor C3. Furthermore, theother ends of the capacitors C1 and C2 and the inductors L1 and L2 arecoupled to a ground terminal.

A high impedance value is applied to the nodes P1 and P2, and this stateis similar to an open state. Furthermore, the relations of C=C1=C2 andL=L1=L2 may be established. Therefore, impedance Za at the node P1 maybe expressed as Equation 1 below.

$\begin{matrix}{{Za} = {j\;\omega\; L\frac{\lbrack {1 - {\omega^{2}{L( {C_{3} + C} )}}} \rbrack}{( {1 - {\omega^{2}{LC}}} )\lbrack {1 - {\omega^{2}{L( {{2C_{3}} + C} )}}} \rbrack}}} & \lbrack {{Equation}\mspace{14mu} 1} \rbrack\end{matrix}$

Here, ω represents an angular frequency (2πf), C represents capacitance,and L represents inductance. Through Equation 1, it can be seen that thecircuit of FIG. 2 has two parallel resonant frequencies and one serialresonant frequency. The inductance values L for deciding the resonantfrequencies are all constant, and the capacitance values C and C3 aredifferent for the respective frequencies. The respective frequencies maybe expressed as Equations 2 to 4 below.

$\begin{matrix}{\omega_{low} = \frac{1}{\sqrt{L( {{2C_{3}} + C} )}}} & \lbrack {{Equation}\mspace{14mu} 2} \rbrack \\{\omega_{mid} = \frac{1}{\sqrt{L( {C_{3} + C} )}}} & \lbrack {{Equation}\mspace{14mu} 3} \rbrack \\{\omega_{high} = \frac{1}{\sqrt{LC}}} & \lbrack {{Equation}\mspace{14mu} 4} \rbrack\end{matrix}$

Here, C represents capacitance, and L represents inductance. As shown inEquations 2 and 4, parallel resonance occurs in the low frequency bandω_(low) and the high frequency band ω_(high), and the impedance Zabecomes infinite. Meanwhile, as shown in Equation 3, serial resonanceoccurs in the middle frequency band ω_(mid), and the impedance Zabecomes zero.

Meanwhile, when an arbitrary input signal Vs is inputted to the node P1,a transfer function for checking a signal V_(P1) at the node P1 may beexpressed as Equation 5 below.

$\begin{matrix}{\frac{V_{P\; 1}}{V_{s}} = \frac{j\;{\frac{\omega\; L}{R_{s}}\lbrack {1 - {\omega^{2}{L( {C_{3} + C} )}}} \rbrack}}{\begin{matrix}{{( {1 - {\omega^{2}{LC}}} )\lbrack {1 - {\omega^{2}{L( {{2C_{3}} + C} )}}} \rbrack} +} \\{j\;{\frac{\omega\; L}{R_{s}}\lbrack {1 - {\omega^{2}{L( {C_{3} + C} )}}} \rbrack}}\end{matrix}}} & \lbrack {{Equation}\mspace{14mu} 5} \rbrack\end{matrix}$

Here, ω represents an angular frequency (2πf), C represents capacitance,L represents inductance, and Rs represents resistance of an arbitraryinput signal Vs. Since the circuit of FIG. 2 is a circuit serving as theload unit 102 b of the LAN 102 in FIG. 1, the resistance Rs may be setto a very large value. The value of the transfer function of Equation 5becomes 1 at the low frequency of Equation 2 and the high frequency ofEquation 4, and the value of the transfer function of Equation 5 becomes1 at the middle frequency of Equation 3.

Similarly, when an arbitrary input signal Vs is applied to the node P2,a transfer function for checking a signal V_(P2) at the node P2 may beexpressed as Equation 6 below.

$\begin{matrix}{\frac{V_{P\; 2}}{V_{s}} = \frac{j\;\frac{\omega^{3}L^{2}C_{3}}{R_{s}}}{\begin{matrix}{{( {1 - {\omega^{2}{LC}}} )\lbrack {1 - {\omega^{2}{L( {{2C_{3}} + C} )}}} \rbrack} +} \\{j\;{\frac{\omega\; L}{R_{s}}\lbrack {1 - {\omega^{2}{L( {C_{3} + C} )}}} \rbrack}}\end{matrix}}} & \lbrack {{Equation}\mspace{14mu} 6} \rbrack\end{matrix}$

Here, ω represents an angular frequency (2πf), C represents capacitance,L represents inductance, and Rs represents resistance of an arbitraryinput signal Vs. The denominator of Equation 6 is equal to that ofEquation 5. This means that the transfer functions of the nodes P1 andP2 are equal to each other in the case of parallel resonance, butdifferent from each other in the case of serial resonance.

FIG. 3 is a graph showing a simulation result for the transfer functionvalues at the nodes P1 and P2 in the circuit diagram of FIG. 2.

In FIG. 3, a solid line indicates the transfer function of Equation 5,i.e., the transfer function at the node P1 which is represented by theunit of dB, and a dotted line indicates the transfer function ofEquation 6, i.e., the transfer function at the node 92 which isrepresented by the unit of dB. Referring to FIG. 3, it can be seen thatthe transfer functions of Equations 5 and 6 have the same value in thecase of the parallel resonance occurring at the low frequency and thehigh frequency, and have different values in the case of the serialresonance occurring at the middle frequency.

As the simulation result, the reason that the transfer function valuesdo not become 1 (0 in the unit of dB) at the low frequency and the highfrequency is that the resistance Rs is set to a very large value. Whenthe resolution of the simulation is significantly increased, thetransfer function values may become 1. For the same reason, the transferfunction value at the middle frequency does not become 0 (negativeinfinity in the unit of dB).

FIG. 4 is a graph showing phases at the nodes P1 and P2 in the circuitdiagram of FIG. 2.

Referring to FIG. 4, a solid line indicates a phase characteristic atthe node P1, and a dotted line indicates a phase characteristic at thenode P2. According to the unique characteristics of an LC resonantcircuit, the phase changes by −180 degrees in the case of the parallelresonance, and the phase changes by 180 degrees in the case of theserial resonance. The phase characteristic at the node P1 is that thephase changes from +90 degrees to −90 degrees at the low frequency ofEquation 2 due to the parallel resonance, and then changes from −90degrees to +90 degrees at the middle frequency of Equation 3 due to theserial resonance. Similarly, the phase changes from −90 degrees to +90degrees at the high frequency of Equation 4 due to the parallelresonance. On the other hand, since the phase characteristic at the nodeP2 does not include serial resonance, the phase characteristic at thenode P2 is that the phase changes by −180 degrees at the low frequencyof Equation 2 and the high frequency of Equation 4, respectively, andfinally becomes −450 degrees.

FIG. 5 is a graph showing a phase difference between the node P1 and thenode P2 in the circuit diagram of FIG. 2.

Referring to FIG. 5, since the serial resonance occurs only at the nodeP1 and does not occur at the node P2, the phase changes by +180 degreesonly at the node P1. Therefore, a phase difference between the node P1and the node P2 corresponds to 180 degrees in a band before the middlefrequency of Equation 3, and 360 degrees in a band after the middlefrequency of Equation 4. That is, a signal having a differential phasecharacteristic of 180 degrees between the nodes at the low frequency ofEquation 2, and having a common phase characteristic of 360 degreesbetween the nodes at the high frequency of Equation 4 is generated.

As such, when the circuit of FIG. 2 is used as the load unit 102 b ofthe LNA 102 in FIG. 1, it is possible to provide a high gain in therespective frequency bands while removing the interference between thefrequency bands in the dual-band wireless communication system.

FIG. 6 is a diagram illustrating the load unit 102 b of the signalconverting apparatus according to a second embodiment of the presentinvention.

Referring to FIG. 6, a varactor Va serving as a variable capacitor iscoupled in parallel to the capacitor C3 in the circuit diagram of FIG.2. The addition of the varactor Va has an effect upon on the lowfrequency of Equation 2 and the middle frequency of Equation 3, but doesnot have an effect upon the high frequency of Equation 4. That is, thevaractor Va may change the parallel resonance of Equation 2 so as toadjust a gain characteristic of the low frequency band between the dualbands to a desired frequency, and may change the serial resonance ofEquation 3 so as to change a frequency which becomes the boundarybetween the frequency bands indicating a differential phase differenceand a common phase difference. Therefore, the phase difference betweenthe low frequency of Equation 2 and the high frequency of Equation 4 isalways maintained at a differential phase and a common phase.

FIG. 7 is a graph showing a simulation result for transfer functionvalues at the node P1 and the node P2 in the circuit diagram of FIG. 6.

Referring to FIG. 7, the simulation was performed on the node P1 and thenode P2, when the varactor Va has a value of 10 fF and when the varactorVa has a value of 500 fF, respectively. As shown in FIG. 7, both whenthe varactor Va has a value of 10 fF and when the varactor Va has avalue of 500 fF, the transfer function at the node P1 and the transferfunction at the node P2 have the same value at the high frequency. Onthe other hand, the transfer function at the node P1 and the transferfunction at the node P2 have different values at the low frequency andat the middle frequency of the serial resonance.

FIG. 8 is a diagram illustrating the load unit 102 b of the signalconverting apparatus according to a third embodiment of the presentinvention.

Referring to FIG. 8, a varactor Va1 is coupled in parallel to thecapacitor C1 and a varactor Va2 is coupled in parallel to the capacitorC2 in the circuit of FIG. 2. The addition of the varactors Va1 and Va2has an effect upon the low frequency of Equation 2, the middle frequencyof Equation 3, and the high frequency of Equation 4. That is, thecircuit diagram of FIG. 8 may control gains in both the low frequencyband and the high frequency band of the dual bands, and the frequencybecoming the boundary between phase differences also changes. Therefore,the condition for removing interference is maintained at all times.

FIG. 9 is a graph showing a simulation result for transfer functionvalues at the node P1 and the node P2 in the circuit diagram of FIG. 8.

Referring to FIG. 9, the simulation was performed on the node P1 and thenode P2, when the varactor Va1 has a value of 10 fF and when thevaractor Va1 has a value of 100 fF, respectively. As shown in FIG. 9,both when the varactor Va1 has a value of 10 fF and when the varactorVa1 has a value of 100 fF, the transfer function at the node P1 and thetransfer function at the node P2 have the same value at the lowfrequency, the middle frequency, and the high frequency.

FIG. 10 illustrates an example in which the circuit diagram of FIG. 2 isapplied to the design of a two-stage LNA.

Referring to FIG. 10, the LNA 1000 includes intermediate tap inductors1020 and 1050, cascode amplifiers 1010 and 1040, and an impedancematching unit 1030.

The impedance matching unit 1030 includes a capacitor C3, an inductorL2, and a transistor M5.

The capacitor C3 is coupled between an input terminal IN and a source ofthe transistor M5, and the inductor L2 is coupled between the source ofthe transistor M5 and a ground terminal. A drain of the transistor M5 iscoupled to a gate of a transistor M1 of the cascode amplifier 1010, anda gate of the transistor M5 is coupled to a power supply terminal forsupplying a bias voltage Vb2.

The transistor M5 is a transistor having a common gate structure, andmay include an amplification element having a control terminal, an inputterminal, and an output terminal. FIG. 10 illustrates that thetransistor M5 is an n-channel field effect transistor (FET).

The capacitor C3 is configured to block a DC voltage at a single-phasesignal VRF inputted to the input terminal IN, and the inductor L2 isconfigured to block an AC voltage at the single-phase signal VRF.

The single-phase signal VRF is inputted to the source of the transistorM5, and then outputted to the gate of the transistor M1 through thedrain of the transistor M5. That is, the single-phase signal VRF may beinputted to the gate of the transistor M1 by the transistor M5, withoutreflection. At this time, a direct current flowing to the source of thetransistor M5 and the size of the transistor M5 may be adjusted toconstantly maintain the input impedance of the transistor M1 at alltimes, regardless of the frequency bands.

Meanwhile, the intermediate tap inductors 1020 and 1050 may be modeledin a similar manner to the circuit diagram of FIG. 2. The intermediatetap inductors 1020 and 1050 include inductors Lc1 and Lc2 coupledsymmetrically and inductors Lc3 and Lc4 coupled symmetrically, andparasitic capacitors between the intermediate tap inductors and thesubstrate symmetrically exist. The intermediate tap inductors and theparasitic capacitors may be modeled into the inductors L1 and L2 and thecapacitors C1 and C2 in the circuit diagram of FIG. 2.

Furthermore, the capacitor C3 in the circuit diagram of FIG. 2 basicallyexists due to such a characteristic that the lines of the intermediatetap inductors are positioned physically close to each other. Therefore,the load unit of each stage in the two-stage LNA according to theembodiment of the present invention serves to simultaneously reducenoise for signals of the dual-band wireless communication system,provide a gain, and remove interference between the respective bands.

The wideband single-phase signal VRF outputted to the drain of thetransistor M5 is converted into a differential phase signal and a commonphase signal by the intermediate tap inductor 1050.

One ends of the respective inductors Lc3 and Lc4 of the intermediate tapinductor 1050 are shared through a tap and then AC-grounded. That is,the one ends of the respective inductors Lc3 and Lc4 may be coupled to apower supply terminal Vdd or ground terminal, or coupled to a powersupply terminal for supplying a specific bias voltage. Here, the oneends of the respective inductors Lc3 and Lc4 should be coupled to aspecific bias voltage for biasing the M1 and M3 transistors.

The other end of the inductor Lc3 is coupled to the input terminal N1 ofthe cascode amplifier 1010 and the drain of the transistor M5, and theother end of the inductor Lc4 is coupled to an input terminal N1′ of thecascode amplifier 1040. The inductor Lc3 serves as a load of thetransistor M5 and causes resonances at two different frequencies.

At this time, the first resonance is caused by the parasitic capacitorcoupled between the inductor Lc3 of the intermediate tap inductor 1050and the drain and source of the transistor M5 and the parasiticcapacitor existing between the inductors Lc3 and Lc4 of the intermediatetap inductor 1050. This corresponds to the low frequency band of thedual-band system. The output signal of the low frequency band isoutputted as a differential phase signal.

The second resonance is caused by the parasitic capacitor coupledbetween the inductor Lc3 of the intermediate tap inductor 1050 and thedrain and source of the transistor M5. This corresponds to the highfrequency band of the dual-band system, and the output signal of thehigh frequency band is outputted as a common phase signal.

Since the resonance frequency is decided by the parasitic capacitors ofthe intermediate tap inductor 1050, the dual bands exhibit a narrow-bandgain characteristic.

In this embodiment of the present invention, the differential phasesignal and the common phase signal which are converted by theintermediate tap inductor 1050 are applied to the two-stage cascodeamplifiers 1010 and 1040, in order to improve the narrow-band gaincharacteristic.

In this embodiment of the present invention, the second amplifier stagehas been proposed in such a manner as to output a signal having adifferential phase as a differential phase signal and to output a signalhaving a common phase as a common phase signal.

Then, the signal of the low frequency band is finally outputted as adifferential phase signal and the signal of the high frequency band isfinally outputted as a common phase signal through the intermediate tapinductor 1020. However, the value of the intermediate tap inductor 1020is set differently from the value of the intermediate tap inductor 1050of the first stage, in order to implement the resonance frequency in aslightly different manner. Through this configuration, the respectivebands of the dual bands may be implemented with a wideband instead of anarrowband.

FIG. 11 is a graph showing gain characteristics and phase differencecharacteristics for the circuit diagram of FIG. 10.

Referring to FIG. 11, the gain characteristic of the high frequency bandbetween the dual bands is not sensitively changed at a resonance pointaccording to an inductance value, unlike the gain characteristic of thelow frequency band. Since a capacitance value multiplied by theinductance value of Equation 2 at the low frequency is considerablylarger than a capacitance value multiplied by the same inductance valueof Equation 4 at the high frequency, the change in the low frequency ofEquation 2 depending on the change of the inductance value is moresensitive than the change in the high frequency of Equation 4.

Meanwhile, both output terminals of the LNA exhibit a phase differenceof 180 degrees at the low frequency and a phase difference of 0 or 360degrees at the high frequency.

FIG. 12 is a graph showing a transient simulation result of thedual-band output signals in the circuit diagram of FIG. 10.

Referring to FIG. 12, it can be seen that a differential phase signal isoutputted at a frequency of 4.5 GHz corresponding to the low frequencyband, and a common phase signal is outputted at a frequency of 10.5 GHzcorresponding to the high frequency band.

According to the embodiments of the present invention, in the wirelesscommunication system receiving single-phase signals, two differentwideband signals having a bandwidth of 1 GHz or more may be received,and reflection and noise figures of the received signals may beminimized to secure isolation between input and output. Then, thesignals of the respective bands may be amplified.

Furthermore, when two different single-phase signals are processed, alow-frequency band signal is outputted as an accurate differential phasesignal, and a high-frequency band signal is outputted as an accuratecommon phase signal. The respective frequency band signals outputted insuch a manner may be transmitted to the next stage without interferencewith each other.

Furthermore, the dual-band receiving apparatus according to theembodiment of the present invention commonly uses the circuit used asthe load unit of the LNA. Therefore, since the receiving apparatus doesnot require an additional amplification stage for acquiring dual bands,the receiving apparatus may have low power consumption and may beimplemented with a small area.

Furthermore, the frequency down-converter including the differentiatorand the combiner may process the dual-band signals transmitted from theLNA. Therefore, the receiving apparatus may have low power consumptionand may be implemented with a small area.

The receiving apparatus according to the embodiment of the presentinvention may be applied to both of a direct-conversion receivingapparatus and a heterodyne-architecture receiving apparatus.

While the present invention has been described with respect to thespecific embodiments, it will be apparent to those skilled in the artthat various changes and modifications may be made without departingfrom the spirit and scope of the invention as defined in the followingclaims.

What is claimed is:
 1. A receiving apparatus in a wireless communicationsystem, comprising: an antenna configured to receive a wirelessfrequency signal comprising a first frequency band signal and a secondfrequency band signal; a low noise amplifier (LNA) configured to amplifythe wireless frequency signal, output the first frequency band signal asa differential phase signal, and output the second frequency band signalas a common phase signal; a differentiator configured to pass only thedifferential phase signal between the signals outputted from the LNA;and a combiner configured to pass only the common phase signal betweenthe signals outputted from the LNA.
 2. The receiving apparatus accordingto claim 1, wherein the first frequency band comprises a lower frequencyband than the second frequency band.
 3. The receiving apparatusaccording to claim 1, wherein the differentiator is implemented as afirst output terminal of a down-converter, and the combiner isimplemented as a second output terminal of the down-converter.
 4. Thereceiving apparatus according to claim 1, wherein the LNA comprises: atransconductance unit configured to perform wideband matching; and aload unit configured to output the differential phase signal and thecommon phase signal.
 5. The receiving apparatus according to claim 4,wherein the load unit comprises a first capacitor, a second capacitor, athird capacitor, a first inductor, and a second inductor, wherein thethird capacitor is coupled between a first node and a second node,wherein one ends of the first inductor and the first capacitor arecoupled between the first node and the third capacitor, wherein thesecond inductor and the second capacitor are coupled between the secondnode and the third capacitor, and wherein the other ends of the firstinductor, the first capacitor, the second inductor, and the secondcapacitor are coupled to a ground terminal.
 6. The receiving apparatusaccording to claim 5, wherein the load unit further comprises avaractor, and the varactor is coupled in parallel to the thirdcapacitor.
 7. The receiving apparatus according to claim 6, wherein thevaractor is configured to adjust a gain of the first frequency band. 8.The receiving apparatus according to claim 5, wherein the load unitfurther comprises a first varactor and a second varactor, wherein thefirst varactor is coupled in parallel to the first capacitor, andwherein the second varactor is coupled in parallel to the secondcapacitor.
 9. The receiving apparatus according to claim 8, wherein thefirst varactor and the second varactor are configured to adjust the gainof the first frequency band and a gain of the second frequency band,respectively.
 10. The receiving apparatus according to claim 1, whereinthe receiving apparatus removes interference between the first frequencyband and the second frequency band using a phase difference between thefirst frequency band and the second frequency band.